Softrock Lite 6.2
Adventures in Electronics and Radio
Elecraft K2 and K3 Transceivers
Diode Turn-on/off Time and Relay
29-30 December 2007; First released
08 April 2009; Revised to add relay snubbing measurements
09 April 2009; Revised to add section "Why Use a Snubber"?
11 April 2009; Added RC snubber section
12 April 2009; Fixed bad graphics link
Table of Contents and page links
Recovery Time Conclusion
This page presents turn-on and turn-off time measurements
for five power diodes:
I took these measurements to answer a debate on the PIC
mailing list concerning the turn-on time of typical silicon power diodes. In
particular, whether a silicon power diode, such as the common 1N400x series, has
an appreciable turn-on delay such that it renders it unsatisfactory to clamp the
inductive kickback of an inductor, such as a relay coil, when switched with a
On 08 April 2009, I've extended the measurements to
include release voltages and operating time for a typical relay without a
snubbing diode and with 1N4148 and 1N4007 snubbing diodes. As my December 2007
measurements demonstrated, a 1N4007 power diode is quite effective as a relay
snubbing device, since the diode's turn-on time is the critical element.
As a preliminary matter, it's well known that standard
silicon power diodes, such as the 1N400x series have a significant turn-off
time, commonly called the "reverse recovery time."
The reverse recovery time involves a diode
switching from forward biased (conducting) to reverse biased (non-conducting).
The diode's PN junction, when conducting, has excess minority carriers and it
requires a finite time for these excess minority carriers to be neutralized when
the PN junction switches from forward to reverse bias. For more detail, see
http://www.microsemi.com/micnotes/302.pdf. Standard silicon power diode,
intended for use in 50/60 Hz power systems, exhibit a reverse recovery time of
several microseconds, although higher performance devices are available.
Less common considered is the forward recovery time,
i.e., the time it takes the diode to conduct when switched from reverse to
forward bias. This mode is involved in the typical inductive clamp circuit, as
can be seen from the circuit fragment below. When the relay coil is energized
(the 2N7000 MOSFET is biased into saturation), diode D8 is reversed biased. When
the 2N7000 MOSFET is turned off, the relay's magnetic field collapses and
induces a reverse voltage in the windings, which forward biases D8. (Lenz's law
says the induced voltage is reversed.)
If D8 has a slow forward recovery time, it is possible for
the voltage induced by the collapsing field to rise to levels that might damage
the switching MOSFET before D8 begins conducting. (This is a simplification, as
the MOSFET will breakdown at some voltage level and will thus limit the voltage.
The question is whether the energy deposited into the MOSFET in this event is
sufficient to damage the device. We'll ignore this aspect of the analysis.)
The circut fragment below shows typical use of a diode to clamp an inductive switching
transient. The time constant is approximately L/R where R is the relay coil
resistance and L is the coil inductance. For an Omron G2RL-24-12DC relay, (12
volt DC coil) the coil resistance is 360 ohms. Omron does not specify the
armature inductance, but based on measurements I've made for similar relays,
it is in the range of 3 H when the armature is closed. L./R is thus 8
milliseconds. The relay's opening will be delayed by a time constant or two,
depending on its release current performance. In addition, of course, there
may be extra delay due to the mechanical structure of the relay armature.
Recovery Time Test Setup
To assess the forward recovery time of typical diodes, I used
the test setup illustrated below.
To see both the forward and reverse recovery time, I set the 8012B to provide a
bi-polar pulse, i.e. from a negative level to positive level. The image below
shows the test signal., with a peak-to-peak voltage of approximately 7.8 volts,
swinging from +3.9V to -3.9V. The rise and fall times are less than 5 ns.
I installed the diodes in a home made test fixture that I use for RF network
analysis. The diode is installed shunting the pulse to ground, such that a
positive pulse puts the diode into forward conduction.
Test fixture with diode shunting the coaxial center
conductor to ground.
is a member of the venerable 1N400x family of silicon rectifiers. I don't know
when they were first introduced, but I recall purchasing some in early 1970's so
they have been around for 35 years plus. Of course, today's 1N400x diode is not
necessarily the same as one fabricated 35 years ago. I have no idea when the
1N4005 I tested was manufactured, as it's been in my junkbox for an unknown
time. (My 1971 GE Transistor/Diode handbook lists the 1N400x series, so the
original introduction date must have been in the mid to late 1960's.)
Fairchild's data sheet for the 1N4005 (the datasheet is
for all 1N400x series) can be viewed at
http://www.fairchildsemi.com/ds/1N/1N4005.pdf. It notes the 1N4005 has a
reverse voltage breakdown of 500 volts, providing no data on either forward or
reverse recovery time. Like all other members of the 1N400x family, the 1N4005
is rated at 1 A forward current.
The horizontal sweep time is 1 μs/div. The vertical spikes
show the applied pulse transition times. Voltage above the center graticule line
are positive, indicating forward bias and those below are negative, where the
diode is reverse biased.
Note how fast the transition from reverse bias to
essentially steady state forward conduction is. The forward bias transition is
way too short to see with the 1 μs/div sweep speed. There's a slight rise in
forward voltage for 200 ns or so after the diode transitions from reverse to
forward bias, but it's only a tenth of a volt or so, negligible in the
application being considered.
The reverse recover time, in contrast, is remarkable. Note
how the 1N4005 continues in forward conduction state essentially unchanged for
400 to 500 ns after the driving waveform switches from positive to negative,
with only a slight drop in voltage. After 500 ns or so, the diode begins to
cease conduction, although the total time to stabilize in reverse mode
approaches 3 μs. Of course, the reverse recovery time is not of
significance in an inductive clamp application.
is the 1000 V reverse voltage member of the 1N400x family, with the same 1 A
forward current specification.
As the illustration below shows, it's not much different
than the 1N4005. The forward recovery time is very short, on the order of a few
nanoseconds, with a similar small transition after switching to forward
The reverse recovery time is similar to the 1N4005.
It's commonly understood that the 1N4007 has an special
junction structure resembling a PIN diode and it has, of course, been widely
used as a "poor man's PIN diode" in RF power switching. Elecraft, for example,
uses 1N4007 diodes for transmit/receive switching in both its 10 watt and 100
watt option K2 transceiver.
Used as a quasi-PIN diode for RF power switching, the
1N4007 takes advantage of the lengthy reverse recovery period, which prevents it
from rectifying the high frequency signal it is switching. (This is because the
reverse voltage half-cycle period of, say, a 3.5 MHz signal is much shorter (140
ns) than the reverse recovery time. Hence, if forward biased with DC, any
reverse bias from RF negative half-cycles don't last long enough to neutralize
the 1N4007's excess minority carriers. Hence, the 1N4007 stays forward biased
and presents a low impedance to the full RF cycle. There's a minimum frequency
for this effect which we see from the trace below. With a bias current in the 60
mA range from the 8012B pulse generator, we see the following voltage waveform.
I measured the forward and reverse current with a Tektronix P6022 clip-on
current probe, set for 1 mA = 1 mV. The figure below shows the diode current.
The figure shows during the forward bias period, the
current is 67.5 mA. Then, when the 8022B pulse generator reverses polarity and
reverse biases the diode, we see the current approximately doubles during the
recombination time. This can be considered as the sum of the recombination
current in the PN junction (68 mA) plus the negative current from the pulse
generator (68 mA) or a total of 136 mA. As the excess minority carriers are
neutralized, the diode assumes its normal reverse bias state of being
essentially non-conducting, with negligible current flow.
Note that since the probe measures current, the forward
turn on voltage elevation appears as a reduced current.
Spehro Pefhany has provided a link to the forward
recovery time graph presented below for the 1N400x series diodes. Unfortunately,
my 68 mA forward test current is just off the horizontal scale, so we have to
extrapolate a bit. Also, the magnitude of the Vf elevation during the forward
recovery time is not provided.
Nonetheless, the forward recovery time graph is in general
agreement with the data I measured.
Let's look at the 1N4007 in more detail with an annotated,
expanded view of the transitions. The illustration is from an 1N4007, but the
concept applies to all diodes.
The transition A'
to A takes the pulse generator goes from negative to positive, and thus the
diode from reverse bias to forward bias. The 'scope sweep shows the transition
to be quite fast, as A'-A appears as an essentially vertical line. With 500
ns/div, we expect to see a transition time that is over 50 ns to show as a
sloping line, so we can say that the 1N4007's turn-on time, or forward recovery
time is < 50 ns. At most, we see a small elevated forward voltage effect for a
couple hundred ns after the diode becomes forward conducting. The diode's
forward voltage starts around 1.2 volts, dropping to the steady state level of
0.8 volts within 250 ns or so.
At time B, the pulse generator output switches from
positive to negative. When negative, the 1N4007 is reverse biased and should
carry no current. Accordingly, we should see on the oscilloscope the full
negative pulse voltage, around -4 volts at time B. However, the 1N4007 continues
in forward conduction at time B, and indeed even by time C, the
diode voltage is still > 0. In other words, during the time from B to C
(approximately 500 ns) the 1N4007 is still in forward conduction mode, despite
the fact that the diode is reverse biased to -4 volts by the pulse generator.
Rather, the diode stays in forward conduction during the time B-C as it takes
time for the excess minority carriers in the 1N4007's PN junction to be
neutralized by the reverse bias.
In fact, it takes until time D for the last excess
minority carriers to be neutralized and the 1N4007 to become fully reverse
biased with essentially no current flow. This is a full 3 μs between the time
the diode is reverse biased and the time that current flow actually ceases.
is the lowest voltage rated of the 1N400x family at 50 volts. Its reverse
recovery switching time behavior is similar to other members of the family, as
evidenced in the curve below.
However, the 1N4001's forward switching shows considerably
less of the increased forward voltage effect seen in the 1N4005 and 1N4007
In fact, if we expand the sweep to 20 ns/div and look at the 1N4001's forward
turn-on time, we see two points of interest:
- The turn-on time is about 4 ns, which represents the
2246's bandwidth (100 MHz) and the rise time of the 8012B pulse generator
(less than 5 ns). The 1N4001's turn-on time is thus less than the test
equipment used in my test setup.
- The excess forward voltage, Vf, is essentially zero.
The overshoot and ringing shown in the 20 ns/div sweep is likely an artifact
of my test setup and cable ringing.
The HER 108
is a member of the HER 10x family, similar to the 1N400x group, with the part
number's last digit increasing with increasing voltage. The HER 10x diodes are
all rated at 1.0 A forward current, and the HEr 108 is rated at 1000 V reverse
voltage. For more detail, see Rectron's data sheet for the family at
However, the HER 10x family is designed to address the
reverse recovery problem seen in the 1N400x and other similar run-of-the-mill
silicon power rectifiers, with a maximum reverse recovery time of 50 ns (for the
HER 101-105 devices, through 400 volts) and 75 ns for the HER 106 - 108 parts
(600 - 1000 volts). No forward recovery time is defined in Rectron's data
The illustration below confirms the major improvement in
reverse recovery time compared with the 1N400x diodes.
The forward recovery time is still short, on the order of
a few nanoseconds, but clear signs of different behavior are seen. The initial
conduction voltage is around 2 volts and it does not drop to the steady state
value of 1 volt for about 500 ns. Both the voltage elevation and response time
in this regard are worse than the 1N400x devices. However, for diode clamping,
the HER 108 seems adequate. Of course, there's no real reason to pay for the HER
108's improved reverse recovery time in this application.
The 11DQ05 is a 1.1 A, 50 volt Schottky diode and more
details may be found at
We expect a Schottky diode to have fast forward and
reverse recovery times, and the 11DQ05 does not disappoint.
To better see the 11DQ05's performance, I've increased the
sweep speed to 0.5 μs/div in the image below. As the illustration shows, both
the forward and reverse recovery times are not easily measurable with this sweep
speed, and are on the order of a few nanoseconds.
Also, the Schottky's lower forward bias voltage drop shows
clearly, with about 0.4 volts across the diode when forward biased.
Recovery Time Conclusion:
a clear difference in reverse recovery time amongst the diodes examined, with
the old 1N400x series having by far the worst reverse recovery time. The HER 108
improved silicon diode provides a much better reverse recovery time, and the
11DQ05 Schottky is even better.
However, for a diode to be used as an inductive snubber,
the evidence is not as clear. All of the diodes transition from reverse to
forward bias with great rapidity. At worse, the HER 108 has a period of higher
than normal forward voltage, but the elevated voltage is not that great when
considered in the context of an inductive snubber.
To one and for all put to rest the myth that a silicon
power diode such as the 1N4007 cannot be used as a relay snubber due to its
recovery time, I set up a simple test fixture, illustrated below.
The HP 8904A multi-synthesizer is configured to provide a
10 Hz square wave (0→5V) driving a 2N7000 MOSFET transistor to turn a
typical relay on and off. DC power is provided by an Agilent E3631A power
supply, set for 13V output. A Tektronix TDS430A digital oscilloscope (400 MHz)
is used to view the circuit's waveforms. The 2N7000 thus turns the relay on for
50 ms and off for 50 ms.
Two measurement setups are used:
- Voltage spike�Channel 1 to test point 1; channel 2 to
test point 2A.
- Relay operate time�Channel 1 to test point 1; channel
2 to test point 2B.
Snubbing Diode Test Setup
The relay used is typical of small footprint relays found in
amateur radio gear. It is a Fujitsu FTR-F1CD-012V part, with a data sheet
available (as of April 2009) at
The key specifications of the relay are 12V nominal operation
(minimum pull-in voltage 9 volts; release voltage 1.2 volts), DPDT ("C-form")
contacts rated at 5A, 24 VDC. Operate time is 15 ms and release time (no diode)
is 5 ms. DC resistance 360 ohms.
The data sheet does not specify the relay coil inductance,
so I measured it with a General Radio GR-1650A RLC bridge, both in the
un-energized and energized mode. (The GR-1650A and B bridges are quite valuable
for measuring relays and capacitors with DC bias, a feature not always available
with lower end digital instruments.)
Computed Stored Energy @
13V, 38 mA
Inductance @ 1000 Hz, un-energized
||828 x 10-6
Inductance @1000 Hz, with DC bias applied to pull in relay
||982 x 10-6
A relay, of course, has different inductance when energized
and non-energized for two reasons. One is mechanical; the magnetic circuit no
longer has an air gap, or if an air gap is designed into the relay, the gap
distance is reduced. This will increase the inductance. Second, the DC bias will
shift the B-H point around which the incremental inductance is measured. In
general the B-H curve shift will usually decrease the inductance.
It would be possible to separate these two effects by
disassembling the relay and mechanically positioning the armature to simulate
energized and non-energized positions whilst measuring the inductance. I'll
leave that as an exercise to the interested reader.
An inductor, including the relay coil, stores energy in
its magnetic field, with the stored energy (in joules or watt-seconds) equal to:
I've computed the stored energy in the above table for the
current the relay consumes at 13V. It's roughly 1 milli-joule when energized.
This stored energy must "go someplace" when the
relay is de-energized. The collapsing magnetic field resulting from opening the
circuit when the 2N7000 goes into a high impedance state, causes a voltage to be
induced in the windings opposite in polarity to the energizing field. (Lenz's
law.) From elementary circuit theory, we know:
E = Ldi/dt
where E is the induced voltage, L is the inductance and di/dt
is the time rate of change of the current. When a fast device, such as a
switching transistor, opens the relay coil circuit in a few hundred nanoseconds,
di/dt becomes quite large.
In the absence of a snubbing diode as shown in the
illustration, the collapsing field can generate a significant voltage,
sufficient to easily cause the 2N7000 to break down. (The 2N7000's rated maximum
drain-to-source voltage is 60 volts.) This completes the circuit and allows the
reverse current flow. Or, if a snubbing diode is present, the diode goes into
forward conduction and allows reverse current to flow through the relay coil. In
either case, the energy stored in the coil's magnetic field is dissipated as
heat in the relay's windings.
This raises an interesting question�what would happen if
we had a perfect switch. Would the collapsing field reach an infinite voltage?
The short answer is no. Our look at the circuit is simplistic, and in fact the
relay windings have distributed capacitance so that an AC current flow can occur
even with one end of the inductor dangling in space, disconnected from the rest
of the circuit. What happens is a dampened oscillation occurs (RLC circuit) with
a resonant frequency determined by the inductance, distributed capacitance and
resistance of the relay coil, with the stored energy dissipated in the coil
resistance over some number of cycles. Although not infinite, it's easy to
generate several hundred volts in this fashion.
with and without Snubber
First, what happens without a snubbing diode? Channel 2
(blue) shows the voltage at the 2N7000's drain; it reaches 68.8 volts and starts
conducting. Since the 2N7000's maximum rated drain-source voltage is 60 volts,
seeing a breakdown at 68.8 volts isn't surprising. It is worth noting that the
2N7000 isn't damaged by this breakdown, or at least the damage is not apparent
in this rather non-critical application. The peak current through the 2N7000 is
limited by the relay's 341 ohm resistance, which for 68.8 volts corresponds to
Next, we add a 1N4148 snubbing diode. The diode limits the peak voltage across
the 2N7000 to 14.4 volts.
Substituting a 1N4007 silicon power diode, we see the same peak voltage, 14.4
volts, across the 2N7000.
Let's look at these waveforms with a faster sweep to see if perhaps the 1N4007
is allowing a glitch or spike too narrow to see with the 500 μs/div
of these oscilloscope captures. We should see a more accurate voltage reading as
well, as whatever happens is rather rapid and 500 μs/div is too slow to
accurately capture the transition.
First, without a
snubbing diode, at 50 ns/division. We see the voltage rises from 0 to 74 volts
in a bit less than 300 ns.
With a 1N4148 snubbing diode, the peak voltage is 13.8 volts and we also see a
damped oscillation with a period of 100 ns, or 10 MHz. If the relay coil
inductance is 570 mH at 10 MHz (which is doubtful at best), for 10 MHz
resonance, the distributed capacitance of the winding is 4.5 x10-16
F. This is clearly an absurd value for a relay coil's distributed capacitance,
so it's likely that the ringing is due to the oscilloscope probe's long ground
lead. (I've written about this before.)
With the 1N4007 diode, the waveform is quite similar to that seen with the
From these measurements, which support the earlier turn-on time data, it is time
to put to rest the mistaken view that a 1N4007 diode cannot be used to snub a
And, there may well be times when a
1N4007's higher current rating is necessary to handle a particular relay. The
small signal relays used inside receivers and transceivers generally have
sufficient coil resistance to limit the voltage spike current to a few hundred
milliamperes, but one should consult the relay data sheet before concluding that
a 1N4148 will work within its ratings.
Why Use a Snubber?
It's not a good idea to allow the relay's Ldi/dt voltage
to cause the switching transistor to break down. Dave Gilbert, AB7E, excellently
summarized the reasons why you should rely upon the switching transistor to
handle the relay inductive spike switching transient in a post on the Elecraft
reflector. Dave has allowed me to quote his posting:
I worked in the discrete semiconductor industry for 30
years (engineering, manufacturing, management) and that statement [that the
switching device breakdown is an acceptable spike limiting methodology] is
simply not true at all.
Zener diodes are designed to break down uniformly
across the entire junction, but most other semiconductors are not. A
reverse voltage breakdown of the collector-base junction will in most cases
force current through an extremely small region of the junction (often
referred to as a "puncture") and fuse the silicon at that point, rendering
the device useless. The current required to do so is much less than
would be required to heat the device even a few degrees, and the time
required to destroy the device is short. Junctions with steeper
diffusion gradients (RF devices, switching transistors, etc) will fail more
easily (sometimes in microseconds), while sloppier junctions (power devices,
etc) will take considerably more abuse. It is not possible to predict
the current at which catastrophic failure will occur.
Emitter-base junctions are typically more graded and
won't fail catastrophically as quickly, but repeated reverse bias conditions
will degrade the transfer gain of the device by creating defect centers that
kill the carrier lifetimes in the base region.
The inductive kick from even a small relay is
sufficient to puncture the junction of many commonly used transistors or
driver ICs, and there are reams of failure analysis reports documenting that
fact. There isn't a manufacturer on this earth that will honor the
warranty for a device used as you describe unless the device was
specifically designed to survive it.
Simulating the circuit with LTspice IV, without a snubbing
diode shows the maximum voltage across the 2N7000 is over 500 volts. This
simulation indicates that the 2N7000 is not well modeled when it comes to
Adding a 1N4148 diode to the simulation shows a response time quite a bit faster
than the measured data. In part this is likely due to the rather slow transition
of the HP 8904A synthesizer which causes the 2N7000 to operate in the linear
region for a few tens of nanoseconds.
Changing the simulation to use a 1N4007 snubbing diode shows virtually identical
Release Time Change
Adding a diode will slow the relay's release/operate
time as the current decay period is longer. It's not possible to show the relay
switching time without a diode because the 2N7000 breaks down and acts as a 70
some volt Zener diode. (Not a good thing for the 2N7000, by the way.) We can measure the times with the three combinations,
however, and compare them with the specification, which is 5 ms in the absence
of a diode. (Yes, I could use a high voltage MOSFET, or, for that matter, a
mercury wetted relay to switch the relay coil. I'll also leave that as an
exercise for the interested reader.)
In the plots below, channel 1 is the 2N7000 gate voltage
and channel 2 is the normally closed contact voltage. (+13 volts is applied to
the common contact.) The time differential between the gate voltage going to
zero and the relay voltage going to +13V represents the relay's release
With no explicit snubber diode (leaving just the 2N7000's
breakdown Zener effect) the release time is 2.68 milliseconds, but note the
contact bounce does not stabilize completely until 8 ms. (The spike at switch
time is due to the inductive voltage from the coil being coupled back into the
oscilloscope probe. The rest of the noise is contact bounce.)
With a 1N4148 snubbing diode, the release time increases to 11.2 ms measured to
the first contact point, or, if allowance for contact bounce is made,
around 18 ms. (The time axis is slowed on the next two plots to 5 ms/div.)
With a 1N4007 snubbing diode, the release time is identical. The
difference in contact bounce is not significant in that bounce varies a bit with
each switching cycle.
Release Time Conclusion
Adding a snubbing diode slows the relay's release time. There
are techniques that reduce the voltage spike with less effect on switching time
but I'll leave that discussion for another time.
Before solid state diodes,
inductive loads were snubbed with RC networks. It's still a useful technique and
worth exploring it allows some control over release time, albeit at the price of
a higher voltage spike. (A Zener diode can be used in a similar fashion, of
The schematic fragment below shows the RC network I designed to snub the
Fujitsu FTR-F1CD-012V relay.
RC Snubber Network for Fujitsu FTR-F1CD-012V Relay
How did I select these particular values? There are several approaches to
designing a snubbing network, and I treated it as a Zobel network. Wikipedia has
an excellent discussion on Zobel networks, which I will not repeat here.
If you are not familiar with a Zobel network, it is one method of canceling
reactance and converting a reactive network to a purely resistive network. The
voltage at the junction of L1-R1 equals the voltage at the junction of R3-C1 for
any frequency. It can be shown that the result is a purely resistive network
when viewed at the network terminals.
relevant equations for the above L-R network are:
R3 = R1
C1 = L1/R12
R3 is simple enough; it equals the relay coil resistance. The nearest 5%
standard value, 330 ohms, can be used for our purposes.
C1 = 0.68 / (341 x 341) = 5.8 x 10-6F or 5.8 μF.
The nearest standard value is 5.6 μF.
with RC Snubber
Let's see how the RC snubber does in the test network. My junkbox did not yield
a 5.6 μF capacitor, so I tried 1, 4.7, 6.8 and 10 μF parts. These should
be non-polarized capacitors, but only the 1 μF I used was non-polarized. I'll
discuss capacitor selection later on this page.
μF & 330 ohm Network
4.7 μF & 330 ohm Network
6.8 μF & 330 ohm Network
10 μF & 330 ohm Network
Although based on a small sample of values, as we move away
from the computed value, the peak voltage across the 2N7000's drain
increases. This is expected as the network becomes reactive (either inductive or
capacitive) as C1 departs from the correct value 5.8 μF. (Complicating things,
the relay's inductance changes as the armature moves, so C1 has no single
"correct" value. Fortunately, this relay presents only a small inductance change
between engaged and released armatures.)
The 4.7 and
6.8 μF parts bracket the computed 5.8 μF value and show the lowest voltage rise,
13.2 volts. Based on voltage rise considerations, any value between 4.7 and 10
μF are acceptable. Even a 1μF capacitor keeps the 2N7000 drain voltage maximum
to around 50% of its rated value.
the RC network alter the relay's release time?
As seen with the diode snubber, there's a price to be paid
for snubbing the inductive spike voltage; slower release time. The same thing is
seen with an RC snubbing network. The test setup is the same as used in the
1 μF & 330 ohm Network
4.7 μF & 330 ohm Network
6.8 μF & 330 ohm Network
10 μF & 330 ohm Network
The 1N4148 diode snubber release time is 11.2 ms. With the
close-to-optimum 4.7 and 6.8 μF capacitor, 330 ohm resistive snubber
network, the release time is 9.6 and 10.3 ms, respectively, a modest improvement
at best. If we allow the 2N7000 switching MOSFET's drain voltage to rise to 30
volts (still only 50% of its maximum 60 volt rating) by using a 1μF capacitor,
the release time improves to 4 ms.
Of course, relay
contact bounce adds additional waiting time in some instances, depending on the
load the relay is switching, but 4 ms is a marked improvement over 11 ms with a
Across the Snubbing Capacitor
Let's look at the voltage across the snubbing capacitor
during the switching interval. I've attached the oscilloscope probes to
points  and  of the snubbing capacitor. Channel 1 is connected to  and
is the black trace in the images. Channel 2 is connected to  and is the blue
trace. I've set both channels to the same gain and the same zero reference
Oscilloscope connections are to  and
. Channel 1 to  and Channel 2 to 
4.7 μF & 330 ohm Network
annotated the oscilloscope capture to identify the polarity reversal area. When
the 2N7000 is pulled low and the relay is enabled and for about 5 ms after the
2N7000 goes high, C1's terminal  is positive with respect to terminal  by
13 volts (during the relay operate period) and a lesser amount after the 2N7000
goes high. The polarity then reverses, with terminal  becoming, at the peak
point, around 4 volts negative with respect to terminal . The polarity
reversal continues for 15 milliseconds or so. The reversal occurs because L1 is
sourcing current into terminal . (I use the positive current convention in
1 μF & 330 ohm Network
sub-optimum 1 μF capacitor, we see a much greater reverse voltage, 15 volts or
so, compared with the 5 volts reverse polarity seen in the close-to-design-value
4.7 μF capacitor.
6.8 μF & 330 ohm Network
With the second close-to-optimum
6.8μF capacitor, the reverse polarity area is roughly similar in duration and
magnitude. (Note the change in horizontal axis sweep time in the next two
plots.) The worst case reverse voltage is around 4 volts.
10 μF & 330 ohm Network
With 10 μF, the reverse voltage
magnitude is even smaller, around 2 volts. As the capacitance increases, the
reverse voltage magnitude will decrease, of course. One can think of this as the
capacitor being able to absorb more current from the inductor without a
corresponding voltage rise. (remember V=q/C where q is charge, C is the
capacitance and V is the voltage. When combined with i = dq/dt where i is the
current and dq/dt is the time rate of change of the charge, it should be
apparent that as C increases, more charge (and hence more input current) can be
held with the same voltage across the capacitor.)
To avoid capacitor failure, therefore, it's necessary to employ a non-polarized
capacitor in the snubbing network. This may be accomplished by using a
non-polarized capacitor or by other methods such as back-to-back polarized
capacitors with parallel diodes to prevent reverse voltage. However, that's a
lot of extra effort to go to in order to avoid buying a non-polarized part.
RC Snubber SPICE
Properly done, computer simulation using SPICE can be
amazingly accurate. As an example, the plot below shows the voltage at points
 =V(junction) and =V(drain) with a 330 ohm / 1 uF RC snubbing network. It
compares quite closely with the measured data presented earlier. The drain peak
voltage, for example measured at 29.6 volts [channel 2 in the oscilloscope
capture] compared with 28.1 SPICE predicted. Likewise the peak voltage at point
 [channel 1 in the oscilloscope capture] measured 23.8 volts peak, compared
with a SPICE predicted peak of 24.9 volts.
The main difference is the time scale; the SPICE
simulated drain voltage oscillation has a period of 5.5 ms, whilst the measured
period is closer to 10 ms, nearly twice as long. I'm not sure what's behind
The SPICE simulation holds the inductance and capacitance
constant. However, the relay coil inductance is not a constant; we know it
varies from 680 mH to 560 mH as the armature releases. In addition, the 1.0 uF
snubbing capacitor is not constant either. The part I used is a high dielectric
constant capacitor normally used for bypassing. As can be seen at
http://www.cliftonlaboratories.com/capacitor_voltage_change.htm these parts
possess a significant C versus V variation. SPICE permits modeling both
non-constant inductance and capacitance, but I have not taken the
simulation to that step.
SPICE Simulation 330 Ohm R and 1 uF C.