Prior Products - no longer available
Softrock Lite 6.2
Adventures in Electronics and Radio
Elecraft K2 and K3 Transceivers
This page presents forward voltage/current measurements on
1N400x and other diodes. Both room temperature and temperature data is
22 January 2008. Added data for two additional diode types; added
temperature/current data for both an 1N4005 diode and the 2SC1945 RF power
transistor. (Click here
to jump to temperature/current data.)
14 April 2009. Added pulse measurement data for 1N4007 and SR2010
Schottky diode. (Click here to jump to
Characteristics of 1N400x Family Diodes
I've been looking at a low power HF power amplifier for
the last week or so, and it's been quite educational in many aspects. One of
many things it has caused me to look at is whether the standard data sheet for a
1N400x diode is accurate when it shows the forward voltage versus current as a
single curve for every member of the family from the lowly 1N4001 through the
Before digging into the diode data, it may be useful to
understand why a standard 50/60 Hz power rectifier diode is relevant to a 7 MHz
solid state amplifier. The answer is to provide a temperature stable operating
point for the amplifier transistor. I can't explain why without at least a few
equations, so please try to follow along.
the bipolar junction transistor as illustrated below.
IC is the collector current and IB.
These are DC values, hence the use of capital letter for the
subscripts, the normal convention when DC values are referenced.
The relationship between the transistor's collector current
and the base current is given by the familiar equation:
α is the common base forward current transfer ratio.
β is the common emitter forward current gain.
ICBO is collector current with the base cut off (connected to the
emitter in the configuration show above. This is also sometimes called the
collector leakage current.
For most of our amateur
radio designs, we work with common emitter circuits, as illustrated above, and
we use β as the design parameter. β is also known as the DC current gain, HFE
and is most often identified that way in data sheets. If a particular transistor
has HFE = 100, and if we want 10 mA collector current, we must inject
100 μA base drive into the transistor.
This deals with the βIB portion of Equation 1.
We then wish to look at the second term, (1+β)ICBO. For a modern
silicon transistor at room temperature, ICBO is a relatively small
number and, even when multiplied by (1+β) can be ignored. For example, the
garden variety small signal NPN transistor, the 2N440 data sheet provides the
If we wish the 2N4401 to have a quiescent current at room temperature of 10 mA,
we set the bias current at 10 mA / 80, or 125 μA. The (1+β)ICBO term
accounts for an additional collector current of (80+1)0.1 μA = 8.1 μA. Compared
with the 10 mA collector current defined by IB, the 8.1 μA leakage
current is negligible.
You might have noticed my
frequent use of the phrase "at room temperature." That's because ICBO
has an exponential increase with temperature. For a silicon transistor, in
theory, ICBO doubles with every 7°C increase in temperature above
25°C, but this is generally pessimistic for most silicon devices.
Let's look at the 2N5109, a good choice for a 1 or 2 watt
In Class A operation, we might bias the 2N5109 at, say, 50 mA.
At this current, the 2N5109 has an HFE of 40 (minimum) to 120
(maximum), so we'll use 80 as the mid-point. At room temperature, the (β+1)ICBO
term is 1.6 mA, which is still relatively small compared with the desired 50 mA
quiescent current. But, when used to deliver 1 or 2 watts output power, the
2N5109's case temperature (and hence junction temperature) will be well above
room temperature. We might not operate it at a case temperature of 150°C, but
operating at 80°C is certainly not out of reason. In this event, the collector
leakage current cannot be ignored and indeed, may be equal to or greater than
the desired quiescent bias current at elevated temperatures. Worse yet, β itself
has a temperature dependency, increasing with increasing temperature.
Worse yet, the transistor may enter thermal runaway, where
the heat generated in the transistor due to the leakage current plus any
intentional quiescent collector current plus signal current increases the
temperature, etc. The result can easily be a destroyed device.
A common fix to prevent thermal runaway is to decrease the
transistor's bias current as the temperature increases. Now we finally get back
to the 1N400x series diodes. At any particular current, the voltage across a
silicon diode such as the 1N400x family, decreases about 2 mV/°C. Hence, if we
derive the amplifier transistor's bias IB from a 1N400x diode that is
thermally connected to the amplifier transistor, the transistor's collector
current tendency to increase with increasing temperature due to leakage current
can be offset by reducing the bias current.
The figure below shows data I gathered a few days ago for
a 2SC1945 RF medium power transistor, suitable for the 5 - 10 watt output level.
The transistor had no RF drive applied, just a constant bias voltage sufficient
to provide a quiescent current of 25 mA at room temperature. Collector voltage
is 13.6 volts in this test. The transistor was mounted on a heat sink. As
the data shows, the 2SC1945's current was heading sharply upward as the the
transistor heated up. Without the power supply's current limiting taking effect,
the 2SC1945 would have been destroyed in a few minutes. Note how rapidly the
current increases in a positive feedback mechanism, where increased temperature
causes increased power dissipation, which in turn increases the temperature,
once the transistor's junction reaches an elevated temperature.
I modified the bias arrangement to use a 1N4005 diode,
thermally linked to the 2SC1945 heat sink. This test also had a brief period of
transmission. Otherwise the test conditions were the same as in the above plot.
As demonstrated, the 1N4005 diode temperature based bias arrangement works well
to prevent thermal runaway. This particular arrangement is not quite right yet,
however, as when the transmit period stops, the quiescent current should
immediately return to the desired rest value, not decaying to that point over
the space of several minutes as shown below.
Finally, we can return to our departure point - the
relationship between forward current and voltage across the 1N400x diodes. This
is important because our thermally linked diode compensation mechanism works the
best when the diode can be set to provide the desired bias voltage to the
transistor when driven by current approximately equal to the power amplifier's
base bias current.
I collected a sample of 5 of the following 1N400x diode
types from my parts supply, as well as other diodes. All 1N400x family members are rated at 1A forward current.
- 1N4001 (50 volts reverse voltage)
- 1N4003 (200 volts reverse voltage)
- 1N4004 (400 volts reverse voltage)
- 1N4005 (600 volts reverse voltage)
- 1N4007 (1KV reverse voltage)
- HER108 (Taiwan Semiconductor "high efficiency
rectifier" 1A forward current, 1KV reverse voltage)
- MJE182 NPN transistor, diode connected
- International Rectifier, 6F80, 6 ampere 800 volt
stud-mount silicon power diode
Fairchild forward characteristics data for 1N400x diode
family. Apparently this curve applies for all 1N400x diodes, from 1N4001
If we look at the physics behind a silicon diode, the
simplest, but still useful, forward voltage versus current relationship is given
by the equation below.
I is the current through
the diode PN junction
IS a reverse saturation component independent of junction potential
e is the charge on an electron
V is the voltage across the diode's PN junction
k is Boltzmann's constant
T is the junction temperature in Kelvin
[e is the base of the natural logarithm, 2.7183...]
Equation 2 says that the relationship between current and
voltage is exponential, and therefore if plotted on a log current axis and a
linear voltage axis, it should be a straight line. This equation also explains
why the diode's voltage drops with increasing temperature, of course.
More importantly for our purposes, the equation has no
terms related to the diode's reverse breakdown voltage, i.e., the exact nature
of the PN junction (doping, etc.) does not appear in equation 2. If we look in
more detail, however, we find that in fact, there is an effect upon the diode's
VI characteristics tied to various PN junction parameters. I won't provide
further detail, but you can learn more about it at
http://materials.usask.ca/samples/DiodeDesign.pdf. Hence, we expect that the
single forward VI chart provided in the data sheets for the entire 1N400x family
is an oversimplification.
And, in fact, Taiwan Semiconductor's HEP10x family
datasheet shows exactly this relationship--the forward voltage drop is related
to the reverse breakdown voltage. (HER108 = 1KV, HER104=300V and HER105=400V.)
In general, for a constant current, the diode's forward voltage is proportional
to the reverse breakdown voltage.
I collected data on my diode samples using an HP6038A
digitally controllable power supply to provide forward current to the diode, and
measured the current through the diode with an Agilent 34410A 6.5 digit
multimeter and measured the voltage across the diode with an HP3456A 6.5 digit
multimeter. The data collection is under software control over an GPIB bus, with
a computer program I wrote.
By applying DC to the diodes under test, the junction
temperature increases, perhaps significantly, at higher current levels. For this
reason, the industry standard is to measure VI with a 300 μs pulse. I don't have
a suitable pulse generator and it would significantly complicate my automated
data collection. The effect of the DC test protocol is to artificially decrease
the forward voltage at higher currents, due to the -2mV/°C diode slope.
My data shows no consistent trend relating breakdown
voltage to forward voltage for my 1N400x samples. I suspect the reason for that
is that my junkbox diodes are of disparate manufacture, both in terms of who
made them and when they were made. I know some of these parts are at least 25
years old, and others are of recent manufacture. During this time, there's been
an evolution of semiconductor processing, even in simple parts such as the
1N400x devices. The data (except for the HER108 parts) plots quite closely to
the straight line Equation 2 predicts.
The data also shows how strange the HER10x parts are,
compared with the 1N400x devices. Apparently Taiwan Semiconductor has something
different in these devices. I also note considerable spread in device-to-device
data for the HER108 parts.
The revised (22 January 2008) data plot shows two
additional diodes. The first is an MJE182 NPN power transistor connected as a
diode. The second is a 6 ampere, stud-mount power rectifier, an International
Rectifier 6F08 part. The 6F08s are old parts, although still in production. I've
had these parts at least 20 years so a modern run of 6F08s might have somewhat
As a 6 ampere rated
diode, the 6F08 is expected to have a larger PN junction area than a 1A rated
diode, which will, in turn, be reflected in lower forward voltage drop at,
e.g., 1A current, compared with a 1N4001, rated at a maximum forward current
of 1A. The data reflects this expectation.
Expanded view of the area of operation most useful for bias
The remainder of this page provides individual family member
plots. The "wiggles" in the data at low current levels results from my operating
the 34410A digital multimeter on a fixed 1A current scale, instead of allowing
the instrument to auto-range. Even a 6.5 digit meter on the 1A scale will have
trouble with currents in the 1μA range, particularly when I did not take the
degree of care required to minimize noise.
intentionally disabled auto-range because as the 34410A changes current ranges,
the burden voltage (current sensing resistor inside the 34410A) changes. (This
is not unique to Agilent's 34410A, I believe.) The changing series resistance
puts bumps in the data as the diode current changes with every range change.
A critical part of temperature compensation of an RF
bipolar transistor amplifier is the slope of the forward voltage versus
temperature curve. In order to characterize the 1N4005 diode, I made a
simple water bath holder by milling out a cavity from a small block of solid
aluminum. The white disk is Teflon. The brown wire parallel with the diode lead
is a Fluke 80PK-1 Type K bead thermocouple. I use a Fluke 189 digital multimeter
to read the water bath temperature with the thermocouple.
The diode is driven with an HP E3610A DC power supply,
operated in constant current mode. The current is measured with a Goldstar
DM7241 digital multimeter and the voltage across the diode is measured with an
HP 3468A digital multimeter. I periodically stirred the water to reduce
temperature gradients. Heat is provided by a small hot plate, with a large
aluminum block to more uniformly distribute the heat source.
Water bath holder.
The large bottom aluminum block helps distribute the hot
plate heat uniformly.
I collected data for three forward current levels, 5, 10 and 35 mA (nominal)
values. The actual currents are slightly different than these nominal values.
Most references simply state that a PN diode junction has
a temperature coefficient of -2 mV/°C. The data I collected, however, confirms
the more detailed PN junction analysis, which says that as the forward current
increases, the temperature dependency slope reduces. My data shows an excellent
fit to the linear relationship, i.e., the diode's forward voltage changes a
certain amount for every degree temperature change, as illustrated by how
closely the straight line regression line matches the data. However, the slope
of this regression fit is not always -2.0 mV/°C.
In fact, SPICE simulation programs do quite well at
predicting this effect. LTspice shows the following, for example.
LTspice circuit. An 1N4005 diode is driven by a constant
current source, with three level steps; 5.1, 10.1 and 34 mA.
The plot below is the LTspice simulation output. The absolute
voltage values differ significantly from my measured data; at 20 °C, for
example, at 10.1 mA bias current, LTspice determines the forward voltage as 614
mV. I measured 678 mV, a considerable difference.
However, the temperature versus voltage relationship for all three bias currents
match my measured data quite closely:
|Diode Forward Current
||LTspice Predicted Slope
Difference from Measured
In all three cases, LTspice shows a steeper slope than
measured, although except for the 5.1mA case, the difference between simulation
and measurement is well within my experimental error.
After all this measured data, how do we determine what diode slope is necessary
to compensate for a particular power transistor? To get a better fix on this
question, I made a series of temperature sweep measurements on an 2SC1945
transistor. Whether this particular transistor is typical or atypical of the
2SC1945 devices, I do not know. And, since I don't have an environmental
chamber, I don't intend to repeat these measurements soon, as they are very time
The data plot presented below show the
effect of increasing temperature on collector current with a fixed bias voltage
level. No signal is applied during this test. The temperature is of the heat
sink to which the 2SC1945 is connected, and the heat sink is warmed with a
combination of collector current (applied voltage 13.6 volts) and warmth from a
hot air gun. The heat sink temperature is monitored by a Fluke 80PK-1 bead
thermocouple and Fluke 189 digital multimeter.
The figure below may seem complicated, but it simply
presents collector current versus temperature for a fixed DC bias voltage
applied to the 2SC1945's base.
I ran the entire family of curves to better understand
what goes on--it would be much easier to run a single constant current sweep.
For example, set the collector current to the desired idle current at 20°C.
Increase the temperature in, say, 20°C steps, reducing the VBE bias voltage to
maintain the target idle current. By taking half-a-dozen similar measurements,
you can characterize the necessary VBE slope rather easily.
If, for example, we decide that the 2SC1945 should be
biased at 10 mA idle current, we can read horizontally across the 10 mA line and
determine that at 20°C, VBE, the base bias voltage, should be about 636 mV. At
80°C, however, VBE must drop to 504 mV. Thus, the VBE source must have a voltage
versus temperature slope of (504 mV - 636 mV) / (80°C -20°C) = -2.2 mV/°C.
If we wish to use a 1N4005 diode to generate the 2SC1945's bias voltage
reference, we need to pick a current that (a) yields 636 mV at 20°C and (b) has
a voltage versus temperature slope of -2.2 mV/°C.
The best match for this to use the 1N4005 characterized in the temperature
run, biased at 5.1 mA. This meets the starting point voltage requirement rather
closely (and, in fact, that's why I picked 5.1 mA as a test parameter), but the
slope is 10% or so short of the required -2.2 mV/°C. Thus, we expect this
particular 2SC1945 transistor biased with this particular 1N4005, set with a 5.1
mA forward current, to exhibit increased collector current with increasing
temperature, although the degree of error may be acceptable.
This configuration is illustrated in the black line below.
In fact, there is a moderate elevation in bias current at elevated temperatures,
as the 2SC1945's target 10 mA idle current increases to around 16 mA over the
full 20-100 degree C range. However, this degree of increase in idle current may
well be acceptable.
To illustrate the importance of getting the diode's
compensation slope to match the required slope, consider the red line in the
plot. It shows the effect of setting the 1N4005's forward current at 35 mA.
Although the idle collector current starts at 10 mA, it soon increases, reaching
unacceptable levels at even modest temperatures.
In this example, it turned out that we could set the 1N4005's
forward current at a level where (a) the diode's forward voltage was exactly the
voltage we needed to establish the 2SC1945's idle current and (b) the diode's
voltage versus temperature slope was reasonably close to -2-2 mV/°C, required to
exactly stabilize the 2SC1945.
In general, it is not
possible to simultaneously meet requirements (a) and (b). Further, even if luck
makes it possible to meet (a) and (b) in a particular case, we almost always
will need to provide more current to the RF transistor than can extracted from
the diode and still meet conditions (a) and (b). These requirements suggest a
more complicated bias circuit than a simple shunt diode can provide. In fact,
the bias circuit used in the test presented above uses one op-amp and current
booster transistor, and decouples bias current generation from bias voltage
reference. I'll write more about this later.
Pulse measurement data
In early 2008 when I measured the diode data, I didn't
have a suitable method for generating a high value constant current drive pulse
and hence used a continuous DC test current to develop the Vf versus If curves.
The problem with DC measurements at high currents is, as I
mentioned, junction heating. A 1N4007 with a forward current of 1 ampere DC
dissipates nearly one watt and the junction temperature is elevated well above
room temperature. As discussed earlier, every 1°C increase in junction
temperature reduces the forward forward voltage by more or less 2 mV.
The 1N400x series diodes has a thermal resistance of
100°C/watt according to Diodes Incorporated datasheet. At 1 ampere DC forward
current, I measured Vf at 0.85 volts, for 850 milliwatts dissipation. With a
25°C room temperature, the test diode's junction temperature was approximately
25°C + 0.85 watts * 100°C/watt or 110°C. (The maximum permitted junction
temperature according to the specification sheet is 150°C.)
More pertinently for the subject of this discussion, the
85°C junction temperature rise reduces the forward voltage by around 170 mV. (As
we've seen, the 2 mV/°C slope is a function of current, but we'll use it as an
estimate.) In the absence of junction heating, therefore, the true Vf would be
1020 mV. How then do we measure the diode's Vf / If characteristics without
contaminating the data with junction temperature rise?
The data sheet provides an answer; the Vf / If data is
collected with a low duty cycle pulse waveform. The norm appears to be a 300μs
pulse with a 2% duty cycle. Assuming the junction has sufficient thermal mass to
average out the pulse heating, a 1 ampere pulse and approximately 1 volt forward
voltage corresponds to a DC power of 20 milliwatts, which translates into a
temperature rise of 2°C and 4 mV temperature error. These values are
sufficiently low that they are negligible for our purpose.
The figure below shows the test setup. The HP8904A
multifunction synthesizer outputs a variable amplitude pulse of 300 microseconds
width. I've set the repetition frequency to 20 pulses per second, with a
corresponding 0.6% duty cycle is.
The 8904A output drives one of two Kepco bipolar power
supply / amplifiers, a BOP 100-1M for up to 1 ampere drive current or a BOP
20-10M for up to 10 ampere drive current. The BOP power supplies are run in
current amplifier mode, so the 8904A's voltage pulse output is translated into a
constant current pulse by the BOP amplifier. (For example, a 1 volt peak pulse
from the 8904A causes the BOP 100-1M to generate a 100 mA constant current
pulse.) The voltage across the diode under test is measured with a Tektronix
TDS-430A digital oscilloscope.
I verified the accuracy of the constant current pulse with
1 ohm and 10 ohm resistors and found it within 2% of the nominal value.
Pulse measurement test setup
The figure below shows both DC and pulse based Vf/If
measurements, as well as data extracted from On Semiconductor's 1N4007 data
As expected, the pulse measured data shows
higher Vf for a given current level once junction heating becomes important. At
1 ampere forward current, for example, the pulse data forward voltage is
elevated by an amount quite close to our 170 mV estimate. My measured data
agrees closely with the data sheet, at least up to two amperes or so. At higher
current levels, the data sheet shows lower forward voltage than I measured. Some
of this difference may result from my choice of 0.6% duty cycle test waveform
compared with the 2% duty cycle employed in the data sheet. At 5 amperes forward
current, a 2% duty cycle test waveform still represents 110 milliwatts
dissipation, which elevates the junction temperature 11°C, causing 22 mV or so
voltage depression. The 0.6% duty cycle test waveform in contrast is 36
milliwatts, so about 15 mV of the difference is attributable to the difference
in junction temperature.
There's also sample-to-sample variation amongst different
diodes and my very simple test fixture is another error source.
I also looked at how a typical Schottky power diode compares
with the venerable 1N4007 silicon rectifier. The SR2010 Schottky diode is rated
at 2 amperes average forward current.
As the data
shows, at low currents the Schottky diode exhibits much lower forward drop than
the silicon part. However, at higher current levels, the difference diminishes.
At 1 ampere and above, for example, the Schottky has between 0.2 and 0.25V less
drop. At 100 mA, in contrast, the Schottky diode has 450 mV less forward drop.